1) Field of the Invention
The present invention relates to switching power supply apparatuses; particularly, relates to apparatuses where a DC magnetic flux component, which is generated in switching power supply apparatuses operating in such a condition that the magnetic flux generated in a transformer is unbalanced, is cancelled with the aid of output current or input current of the power supply apparatus, so that the efficiency of the apparatus is improved.
2) Related Art Statement
Such a switching power supply apparatus where a direct magnetic flux component generated in switching transformers is cancelled is disclosed in U.S. Pat. No. 6,304,460.
FIG. 30 shows an example of the power supply apparatus disclosed in U.S. Pat. No. 6,304,460. The circuit shown in FIG. 30 corresponds to that in FIG. 35b of U.S. Pat. No. 6,304,460. In order to make the following explanation easy the position of the windings are modified in FIG. 30, however, the circuits in FIG. 30 and FIG. 35b of U.S. Pat. No. 6,304,460 are electrically equivalent.
The basic operation of the circuit shown in FIG. 30 will be explained. The power supply apparatus comprises a DC power source 100, and a transformer 101; a main switch 103, a sub-switch 108, a bypass capacitor 102 and a clamping capacitor 109 are provided on the primary side of the transformer 101; rectifying switches 104 and 107, a smoothing capacitor 105 are provided on the secondary side to output a power to a load 106. The transformer 101 comprises a primary winding 101a and a secondary winding 102e, to which an input winding 101b and a magnetic flux canceling winding 101f are further coupled. The apparatus is arranged as a fryback converter where the primary winding 101a and the secondary winding 101e conduct an ON-OFF operation; the main switch 103 and the sub switch 108 are alternatively made ON with a short dead time so as to conduct a so-called active clamping, so that a soft switching operation (zero-voltage switching) is realized. The clamping capacitor 109 is for a voltage clamping; the bypass capacitor 102 is for keeping the plus side of the primary winding 101a at an AC earth voltage. The output from the DC power source 100 is supplied to the transformer 101 via the input winding 101b. 
The electric current coming from the secondary winding 101e is rectified with the rectifying switch 104; then outputted to the smoothing capacitor 105 and the load resistance 106 via the magnetic flux canceling winding 101f. The inductance 101c is a leakage inductance of the input winding 101b or an external inductance, and the inductance 101g is a leakage inductance of the magnetic flux canceling winding 101f or an external inductance.
The rectifying switch 107 is arranged to become ON when the rectifying switch 104 is OFF, so that an electric current is continuously supplied to the smoothing capacitor 105 and the load resistance 106. More strictly, a gate signal is given to the rectifying switch 107 at a timing, which is a little earlier than the timing when the body diode of the rectifying switch 107 is made ON, so that a perfect ZVS (zero voltage switching) operation can be realized in the main switch 103.
In the conventional apparatus, the input winding 101b, the primary winding 101a, and the secondary winding 101e are wound so that the magnetic fluxes generated by these windings have the same direction, but the magnetic flux canceling winding 101f is provided to generate a magnetic flux having a direction that is opposite to the direction of the magnetic fluxes generated by the windings 101b, 101a and 101e. Therefore, by selecting an appropriate number of turns of the winding 101f, it is possible to arrange such that a DC magnetic flux component is not apt to be generated in the transformer 101. When the generation of the DC magnetic flux component is reduced, the gap of the core of the transformer can be made very thin or the gap is not necessary any more, therefore a high permeability can be obtained. Further, since the transformer operates in a working area having less iron loss as well, the number of turns of each winding can be made fewer; both a copper loss and an iron loss can be reduced as a result.
In the conventional apparatus, when the numbers of turn of the primary winding 101a and the input winding 101b are substantially the same, the high frequency current generated by the switching operation is not apt to flow to the input winding 101b and the DC power source 100, so that an EMI noise can be reduced. That is to say, the electric potential of the plus side of the input winding 101b becomes zero in an alternative manner and no potential difference is generated in the leakage inductance 101c, so that the high frequency current is not apt to flow.
At the secondary side of the transformer, when the duty ratio is 0.5, it is possible to prevent that the high frequency current goes to the magnetic flux canceling winding 101f by an arrangement that the number of turns of the magnetic flux canceling winding 101f is equivalently about half of that of the secondary winding 101e. (See, FIG. 33, U.S. Pat. No. 6,304,460). The term “equivalently half” means that, for instance, even when the number of turns of both the windings 101e and 101f is four (4) turns, the equivalent number of turns of the magnetic flux canceling winding 101f is only 2 turn by an arrangement that 50% of the magnetic flux going through the secondary winding 101e goes through in the magnetic flux canceling winding 101f. In the explanation below, the number of turns of the magnetic flux canceling winding means the “equivalent turn number”.
From the point of view of magnetic saturation, when the duty ratio is 0.5, the best operating point where the magnetic saturation is not apt to be occurred, can be obtained by making the number of turns of the secondary winding 101e and the magnetic flux canceling winding 101f equivalently the same. Therefore, the number of turns of these windings may be determined as occasion demands, that the high frequency current is reduced in order to prevent the output ripple, or that the DC magnetic flux is reduced in order to limit the iron loss and the copper loss. It may be also possible to select a compromised number between these two occasions.
The leakage inductance 101c and 101g give a better effect to reduce the high frequency current. The detail of the transformer construction, that makes the leakage inductance larger, is disclosed in U.S. Pat. No. 6,304,460. However, since this is not actually related to the subject of the present invention, the explanation is omitted here. Briefly speaking, by adding an independent winding, that is not connected to the other windings, to an objected winding, where the increased leakage inductance is required. (See, reference LL, in FIGS. 32, 33, 39b, U.S. Pat. No. 6,304,460)
As mentioned above, the apparatus disclosed in U.S. Pat. No. 6,304,460 or the modified apparatus shown in FIG. 30 works in an appropriate manner by making a balance in the number of turns of each winding on the transformer and designing the magnetic circuit and the switching timings in an appropriate manner. That is to say, in the apparatus, the DC magnetic flux is cancelled and the main switch realize a perfect ZVS.